Charge pump with temporally-varying adiabaticity

ABSTRACT

Operation of a charge pump is controlled to optimize power conversion efficiency by using an adiabatic mode with some operating characteristics and a non-adiabatic mode with other characteristics. The control is implemented by controlling a configurable circuit at the output of the charge pump.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. application Ser. No.14/719,815, filed on May 22, 2015, which claims the benefit of thepriority date of U.S. application Ser. No. 14/027,716, filed on Sep. 16,2013, issued as U.S. Pat. No. 9,041,459 on May 26, 2015, the contents ofwhich are hereby incorporated by reference in their entirety

BACKGROUND

This invention relates to adiabatic power conversion, and in particularto configuration and control for partial adiabatic operation of a chargepump.

Various configurations of charge pumps, including Series-Parallel andDickson configurations, rely on alternating configurations of switchelements to propagate charge and transfer energy between the terminalsof the charge pump. Energy losses associated with charge propagationdetermine the efficiency of the converter.

Referring to FIG. 1, a single-phase Dickson charge pump 100 isillustrated in a step-down mode coupled to a low-voltage load 110 and ahigh-voltage source 190. In the illustrated configuration, generally thelow-voltage load 110 is driven (on average) by a voltage that is ⅕ timesthe voltage provided by the source and a current that is five times thecurrent provided by the high-voltage source 190. The pump is driven inalternating states, referred to as state one and state two, such thatthe switches illustrated in FIG. 1 are closed in the indicated states.In general, the duration of each state is half of a cycle time T and thecorresponding switching frequency of the charge pump 100 is equal to theinverse of the cycle time T.

FIGS. 2A-B illustrate the equivalent circuit in each of states two andstate one, respectively, illustrating each closed switch as anequivalent resistance R. Capacitors C1 through C4 have a capacitance C.In a first conventional operation of the charge pump 100, thehigh-voltage source 190 is a voltage source, for example, a twenty-fivevolt source, such that the low-voltage load 100 is driven by five volts.In operation, the voltage across the capacitors C1 through C4 areapproximately five volts, ten volts, fifteen volts, and twenty volts,respectively.

One cause of energy loss in the charge pump 100 relates the resistivelosses through the switches (i.e., through the resistors R in FIGS.2A-B). Referring to FIG. 2A, during state two, charge transfers from thecapacitor C2 to the capacitor C1 and from the capacitor C4 to thecapacitor C1. The voltages on these pairs of capacitors equilibrateassuming that the cycle time T is sufficiently greater than the timeconstant of the circuit (e.g., that the resistances R are sufficientlysmall. Generally, the resistive energy losses in this equilibration areproportional to the time average of the square of the current passingbetween the capacitors and therefore passing to the low-voltage load110. Similarly, during state one, the capacitors C3 and C2 equilibrate,the capacitor C4 charges, and the capacitor C1 discharges, alsogenerally resulting in a resistive energy loss that is proportional tothe time average of the square of the current passing to the low-voltageload 110.

For a particular average current passing to the load 110, assuming thatthe load presents an approximately constant voltage, it can be shownthan the resistive energy loss decreases as the cycle time T is reduced(i.e., switching frequency is increased). This can generally beunderstood by considering the impact of dividing the cycle time byone-half, which generally reduces the peak currents in the equilibrationby one half, and thereby approximately reduces the resistive energy lossto one quarter. So the resistive energy loss is approximately inverselyproportional to the square of the switching frequency.

However, another source of energy loss relates to capacitive losses inthe switches, such that energy loss grows with the switching frequency.Generally, a fixed amount of charge is lost with each cycle transition,which can be considered to form a current that is proportional to theswitching frequency. So this capacitive energy loss is approximatelyproportional to the square of the switching frequency.

Therefore, with a voltage source and load there an optimal switchingfrequency that minimizes the sum of the resistive and capacitive energylosses, respectively reduced with increased frequency and increased withincreased frequency.

SUMMARY

Patent Publication WO 2012/151466, published on Nov. 8, 2012, describesconfigurations in which the source and/or load comprise regulatingcircuits. In particular, in FIGS. 1 and 2A-B, the load 110 caneffectively comprise a current sink rather than present a constantvoltage in an example of what is referred to as “adiabatic” operation ofa charge pump. If the current sink accepts constant current, then thecurrents illustrated in FIG. 2A effectively remain substantiallyconstant values during the illustrated state. Therefore, the resistivepower loss is lower than the resistive loss in the voltage driven casediscussed in the Background, and also substantially independent of thecycle time T. In situations in which the load sinks a pulsed current,then for a particular average current, the resistive energy lossgenerally increases as the duty cycle of the current decreases (and thepeak current increases). There is a range of low duty cycles in whichthe resistive losses with a pulsed current exceed the losses for thesame average current that would result from the charge pump driving arelatively constant output voltage, for example, across a large outputcapacitor.

In one aspect, in general, operation of a charge pump is controlled tooptimize power conversion efficiency by using an adiabatic mode withsome operating characteristics and a non-adiabatic mode with othercharacteristics. The control is implemented by controlling aconfigurable circuit at the output of the charge pump.

In another aspect, in general, operation of a charge pump is controlledso that resistive power losses are minimized by using an adiabatic modewith relatively high duty cycle (i.e., relatively high output current)and using a non-adiabatic mode with relative low duty cycle (e.g.,relatively low output current). In some examples, mode is selected byselectively introducing a compensation capacitor at the output of thecharge pump to present a substantially constant voltage.

In another aspect, in general, an apparatus a charge pump and acontroller coupled to the charge pump. The charge pump has a pluralityof switch elements arranged to operate in a plurality cycles, with eachcycle being associated with a different configuration of the switchelements. The switch elements are configured to provide charging anddischarging paths for a plurality of capacitive elements. The controllerhas an output for controlling timing of the cycles of the charge pumpand one or more sensor inputs for accepting sensor signals characteringoperation of the charge pump and/or operation of peripheral circuitscoupled to the charge pump. The controller is configured adjust thetiming of the cycles of the charge pump according variation of the oneor more sensor inputs within cycles of operation of the charge pump.

In another aspect, in general, an apparatus includes a switchedcapacitor charge pump configured to provide a voltage conversion betweenterminals including a high voltage terminal and a low voltage terminal.The apparatus also includes a compensation circuit coupled to a firstterminal of the charge pump for driving a load by the charge pump, thecompensation circuit providing a capacitance configurably couplable tothe first terminal of the charge pump. A controller is coupled to chargepump and the configurable circuit, and has an output for configuring thecompensation circuit, and one or more sensor inputs for accepting sensorsignals charactering operation of the charge pump and/or operation ofperipheral circuits coupled to the charge pump. The controller isconfigured to configure the compensation circuit according to the sensorsignals to affect efficiency of power conversion between a power sourcecoupled to the charge pump and the load coupled to the charge pump viathe configurable circuit.

Aspects may include one or more of the following features.

The controller is configured to couple a selected capacitance to thefirst terminal to optimize an efficiency of the power conversion.

The one or more sensor signals include a sensor signal thatcharacterizes time variation of a current passing to or from the chargepump via the compensation circuit. In some examples, the sensor signalcharacterizes a duty cycle of a pulsed current passing to or from thecharge pump. In some examples, this current passing to or from thecharge pump via the compensation circuit is a current passing betweenthe compensation circuit and a peripheral coupled to the charge pump viathe compensation circuit.

The one or more sensor signals include a sensor signal thatcharacterizes a voltage at at least one of the terminals of the chargepump and at the peripheral circuit coupled to the charge pump.

The one or more sensor signals include a sensor signal thatcharacterizes switching frequency of the charge pump.

The controller is configured to determine an operating mode from thesensor signals, and to determine the configuration of the compensationcircuit according to the determined mode.

The controller is configured to identify at least a mode having fastswitching limit operation of the charge pump and a pulsed current load,and increase the capacitance coupled to the first terminal in said mode.

The controller is configured to identify at least a mode having slowswitching limit operation of the charge pump and a pulsed current loadwith a duty cycle less than a threshold duty cycle, and increase thecapacitance coupled to the first terminal in said mode.

The apparatus further includes a peripheral circuit that includes aregulator coupled to the compensation circuit. The regulator provides acurrent-based load via the compensation circuit to charge pump. Thecontroller is configured to determining a configuration of thecompensation circuit according to an efficiency of power conversionperformed by the charge pump. In some examples, the regulator comprisesa Buck converter. In some examples, the charge pump comprises aSeries-Parallel charge pump. In some examples, the charge pump comprisesa Dickson charge pump.

In another aspect, in general, a method is directed to power regulationusing a charge pump coupled to a load using a compensation circuitcoupled to a terminal of the charge pump. The method includesconfiguring a capacitance provided by the compensation circuit to afirst terminal of the charge pump. The capacitance is selected accordingto the sensor signals to affect efficiency of power conversion between apower source coupled to the charge pump and the load coupled to thecharge pump via the configurable circuit.

The method may include acquiring the sensor signals. The sensor signalsmay characterize one or more a time variation of a current passing to orfrom the charge pump via the compensation circuit, a duty cycle of acurrent passing between the compensation circuit and a peripheralcircuit, a voltage at the first terminal of the charge pump, and avoltage at the peripheral circuit coupled to the charge pump.

One advantage of one or more embodiments is that efficient operation ismaintained in varying operating modes of the power converter.

Another advantage of one or more embodiments is that a controller doesnot have to be preconfigured for a particular use of a charge pump andcan adapt to the circuit in which the pump is embedded without furtherconfiguration. For example, the controller can adapt to the size of pumpcapacitors used, type of regulator coupled to the pump, switchingfrequency of the pump and/or regulator, etc.

Other features and advantages of the invention are apparent from thefollowing description, and from the claims.

DESCRIPTION OF DRAWINGS

FIG. 1 is a single-phase 1:5 Dickson charge pump;

FIGS. 2A-B are equivalent circuits of the charge pump of FIG. 1 in twostates of operation;

FIGS. 3 and 4 are circuits having a switchable compensating circuitcoupled to the charge pump;

FIG. 5 is a circuit for measuring a charge pump current;

FIG. 6 is a schematic illustrating charge transfer during one cycle ofthe charge pump illustrated in FIG. 4;

FIGS. 7A-C are graphs of output voltage of the charge pump illustratedin FIG. 4 at different output current and switching frequencyconditions; and

FIG. 8 is a single-phase series-parallel charge pump.

DESCRIPTION

As introduced above, as one example, a charge pump 100 illustrated inFIG. 1 may be operated in an “adiabatic” mode in which one or both of alow-voltage peripheral 110 and a high-voltage peripheral 190 maycomprise a current source. For example, Patent Publication WO2012/151466, published on Nov. 8, 2012, and incorporated herein byreference, describes configurations in which the source and/or loadcomprise regulating circuits. In particular, in FIGS. 1 and 2A-B, thelow-voltage load 110 can effectively comprise a current source ratherthan a voltage source in an example of what is referred to as“adiabatic” operation of a charge pump. If the current source maintainsa constant current from the charge pump, then currents illustrated inFIG. 2A maintain substantially constant values during the illustratedstate. Therefore, the resistive losses in the switches through which thecurrent passes are lower than the resistive loss in the voltage loadcase, and also substantially independent of the switching frequency andthe cycle time T. As in the voltage driven case, there capacitive lossesin the switches grow with increasing switching frequency, which suggeststhat lowering the switching frequency is desirable. However, otherfactors, which may depend on internal aspects of the charge pump,voltage or current characteristics at the terminals of the charge pump,and/or internal aspects of the peripheral elements, such as the sourceand/or load, may limit the cycle time (e.g., impose a lower limit on theswitching frequency).

Referring to FIG. 3, in a first mode of operation, a load 320 can beconsidered to comprise a constant current source 312 with an outputcurrent IO. In some implementations, the load 320 also includes anoutput capacitor, which for the analysis below can be considered to besmall enough such that current passing to the load 320 can be consideredto be substantially constant. As introduced above with reference toFIGS. 2A-B, the charge transfer between capacitors in the charge pump100 during the alternating states of operation of the charge pump 100are therefore substantially constant in the adiabatic mode of operation.

Continuing to refer to FIG. 3, a compensation circuit 340 is introducedbetween the charge pump 100 and the load 320. A switch 344 iscontrollable to selectively introduce a compensation capacitor 342 tothe output of the charge pump 100.

Various factors can affect the efficiency of the power conversionillustrated in FIG. 3, including the voltage of an input voltage source392, the switching frequency of the charge pump 100, and the outputcurrent 10 (or somewhat equivalently the input or output current of thecharge pump 100). The efficiency is also dependent on whether or not thecompensation capacitor 342 is coupled to the output path via the switch344. As a general approach, a controller 350 accepts inputs thatcharacterize one or more factors that affect efficiency and outputs acontrol signal that sets the state of the switch 344 according towhether efficiency is expected to be improved introducing thecompensation capacitor versus not. A further discussion of logicimplemented by the controller 350 is provided later in this Description.

Referring to FIG. 4, in another example, a configuration of a chargepump 100 has a regulator 320 coupled via a compensation circuit 340 tothe low-voltage terminal of a charge pump 100, and a voltage source 392coupled to the high-voltage terminal of the charge pump 100. Theregulator 320 (also referred to below generally interchangeably as a“converter”) illustrated in FIG. 4 is a Buck converter, which consistsof switches 322, 324, an inductor 326, and an output capacitor 328. Theswitches open and close (i.e., present high and low impedance,respectively) in alternating states, such that the switch 322 is openwhen then the switch 324 is closed, and the switch 322 is closed whenthe switch 324 is open. These switches operate at a frequency than canbe lower, higher, or equal to the switches in the charge pump 100, witha duty cycle defined as the fraction of time that the switch 322 in theregulator 320 is closed. A preferred embodiment is when the switchingfrequency of the charge pump 100 is lower than the regulator 320.However, in the case the charge pump 100 is at a higher frequency thanthe regulator 320, the charge-pump 100 is disabled when the regulator320 is off (low duty cycle) and the charge-pump 100 is enabled when theregulator 320 is on.

In general, the regulator 320 operates at its highest power efficiencywhen it operates at its highest duty cycle. In some examples, acontroller of the regulator (not shown) adjusts the duty cycle in aconventional manner to achieve a desired output voltage VO. During thecycles of the regulator 320 in which the switch 322 is closed, thecurrent passing from the charge pump 100 to the regulator 320 iseffectively constant, equal to the current through the inductor 326.Assuming that the switching frequency of the regulator 320 issubstantially higher than the switching frequency of the charge pump100, the charge pump 100 can be considered to be driven by a pulsedcurrent source with an average current equal to the duty cycle times theinductor current.

Note that as introduced above, in situations in which the regulator 320sinks a pulsed current, then for a particular average current, theresistive energy loss generally increases as the duty cycle of thecurrent decreases, approximately inversely with the duty cycle. There isa range of low duty cycles, and thereby high peak current relative tothe average current, in which the resistive losses with a pulsed currentexceed the losses for the same average current that would result fromthe charge pump 100 driving a relatively constant output voltage, forexample, across a large output capacitor. Therefore, for a selectedrange of low duty cycles, the controller 350 closes the switch 344 andintroduces a relatively large compensation capacitor 342 at the outputof the charge pump 100. The result is that the charge pump 100 ispresented with a substantially constant voltage, and therefore operatesin a substantially “non-adiabatic” mode. Therefore, the controller 350is effectively responsive to the output voltage because the duty cycleis approximately proportional to the output voltage. Thereby operatingthe charge pump 100 in an adiabatic mode at high output voltage and in anon-adiabatic mode at low output voltage; and switches between theadiabatic and non-adiabatic modes at a threshold duty cycle to maintainan optimum efficiency of the overall power conversion.

Examples of control logic implemented in the controller 350 inconfigurations such as those illustrated in FIGS. 4 and 5 can be underin view of the following discussion.

In general, a charge pump can operate in one of two unique operatingconditions, or in the region in between them. In a slow switching limit(SSL) regime the capacitor currents in the charge pump have the time tosettle to their final values and capacitor voltages experiencesignificant change in magnitude from beginning to end of a cycle of thecharge pump operation. In the fast switching limit (FSL) regime, thecapacitors do not reach equilibrium during a cycle of the charge pumpoperation, for instance, due to a combination of one or more of highcapacitances, high switching frequency, and high switch resistances.

Another factor relates to the capacitance at the output of the chargepump 100, which in the circuits of FIG. 4 can be increased by closingthe switch 344 to add the compensation capacitor 342 to the output. Forsmall output capacitance, the output current of the charge pump 100 iseffectively set by the pulsed current characteristic of the regulator320. As discussed above, for a given average current, the resistivepower losses in the pulsed current case are approximately inverselyproportional the duty cycle.

For large output capacitance, the RMS of the output current of thecharge pump 100 is effectively determined by the equilibration of theinternal capacitors of the charge pump 100 with the compensationcapacitor 342 and the regulator 320. For a given average current, thisresistive power loss is approximately inversely proportional to thesquare of the peak-to-peak voltage across the internal capacitors in thecharge pump 100.

Four combinations of FSL/SSL and constant/pulsed IO modes of operationare possible. In some examples, each of these four modes is affected indifferent ways based on the addition of a compensation capacitor 342 asshown in FIGS. 3 and 4.

Case one: In FSL mode, with constant output current IO as in FIG. 3,introduction of the compensation capacitor 342 does not substantiallyaffect conversion efficiency.

Case two: In FSL mode with pulsed output current as in FIG. 4,efficiency increases when the compensation capacitor 342 is introduced,thereby reducing the RMS current seen by the charge pump 100.

Case three: In SSL mode, with constant output current IO as in FIG. 3,efficiency generally increases without introduction of the compensationcapacitor 342, thereby yielding adiabatic operation.

Case four: In SSL mode, with pulsed load current as in FIG. 4,efficiency depends on the relation between the average output current,the duty cycle, and how far the charge pump 100 is operating from theSSL/FSL boundary. For example, at low duty cycle, efficiency generallyincreases with introduction of the compensation capacitor 342, therebyyielding non-adiabatic operation. In contrast, at high duty cycle,efficiency generally increases without introduction of the compensationcapacitor 342, thereby yielding adiabatic operation. Furthermore, whenthe charge pump 100 is in SSL mode, the farther from the SSL/FSLboundary, the lower the duty cycle at which the efficiency trendreverses.

Depending on the relative values of charge pump capacitors, switchresistances and frequency, it is possible that the charge pump operatein a regime between FSL and SSL. In this case, there is effectively atransition point between case four and case two at which thecompensation capacitor is introduced according to the overall efficiencyof the conversion. As described above, knowledge of the average chargingcurrent and its duty cycle is necessary in case four for determining ifintroduction of the compensation capacitor will improve efficiency.

In some implementations, the controller 350 does not have access tosignals or data that directly provide the mode in which the powerconversion is operating. One approach is for the controller to receive asensor signal that represents the input current of the charge pump, andinfer the operating mode from that sensor signal.

As an example, a sensor signal determined as a voltage across the switchat the high voltage terminal of the converter (e.g., the switch betweensource 109 and the capacitor C4 in FIG. 1) can be used to represent thecurrent because when the switch is closed, the voltage is the currenttimes the switch resistance.

An alternative circuit shown in FIG. 5 provides a scaled version of theinput current IIN. The input switch 510, with closed resistance R is putin parallel with a second switch with closed resistance kR, for example,fabricated as a CMOS switch where the factor k depends on the geometryof the switch. When the switches are closed the differential amplifier530 controls the gate voltage of a transistor 540 such that the voltagedrop across the two switches are equal, thereby yielding the scaledinput current IIN/k, which can be used to form a sensor input signal forthe controller.

The sensed input current can be used to determine whether thecompensation capacitor should be switched in, for example, according toa transition between case four and case two described above.

One possible method for determining the operation mode of the chargepump 100 consists of taking two or more measurements of the inputcurrent IIN and establishing that the difference between the values ofconsecutive samples is substantially zero for SSL mode, or is above apre-determined threshold for FSL mode.

Another method is to measure the difference in the voltage of acapacitor in the charge pump 100. Once the input current IIN is known,the controller 350 can infer the operating mode based upon the voltageripple on the capacitor over a full cycle. Note that the controller 350does not necessarily know the particular sizes of capacitors that areused in the charge pump 100, for example, because the capacitors arediscrete capacitors that are not predetermined. However, the capacitorvalues can be inferred from knowledge of the current, voltage ripple,and frequency, thereby allowing the controller 350 to determine whetherthe charge pump 100 is operating in the FSL or SSL mode. The controller350 can then select adiabatic or non-adiabatic charging by controllingthe switch 344 to selectively introduce the compensation capacitor 342.

Other controller logic is used in other implementations. For example, analternative is for the controller to measure efficiency given by:

η=VO/(N*VIN)

where η is the efficiency, VO is the measured converter output voltage,VIN is the measured converter input voltage, and N is the charge pumpconversion ratio.

The controller directly measures the effect of selecting adiabatic vs.non-adiabatic charging on converter efficiency by comparing the averagevalue of the output voltage VO over a complete charge pump cycle.

Other controller logic uses combinations of the approaches describedabove. For instance, the controller can confirm that the assessment ofcharge pump operating mode and estimation of efficiency increase bychanging the charge pump charging mode.

A traditional method for operating the charge pump 100 is at a fixedfrequency in which the switching occurs independently of the loadrequirement (i.e., the switches in FIG. 1 operate on a fixed timeperiod). Referring to FIG. 6, during one cycle of the switching of thecharge pump 100, a current I1 discharges from the capacitor C1 and acurrent IP discharges other of the capacitors in the charge pump 100.For a particular intermediate current IX, the longer the cycle time T,the larger the drop in voltage provided by the capacitor C1. Aconsequence of this is that the switching frequency generally limits themaximum intermediate current IX because the switching frequency for aparticular load determines the extent of voltage excursions, and in somecases current excursions (i.e., deviations, variation), at variouspoints and between various points within the charge pump 100 and at itsterminals. For a particular design of charge pump 100, orcharacteristics of load and/or source of the charge pump 100, there areoperational limits on the excursions.

Referring to FIGS. 7A-C, the intermediate voltage VX of the charge pump100 is shown in various current and timing examples. Referring to FIG.7A, at a particular intermediate current IX, the intermediate voltage VXgenerally follows a saw-tooth pattern such that it increases rapidly atthe start of each state, and then generally falls at a constant rate.Consequently, the rate of voltage drop depends on the output current IO.At a particular output current IO and switching time, a total ripplevoltage δ results, and a margin over the output voltage VO ismaintained, as illustrated in FIG. 7A. (Note that the graphs shown inFIGS. 7A-B do not necessarily show certain features, including certaintransients at the state transition times, and related to the highfrequency switching of the regulator 320; however these approximationsare sufficient for the discussion below).

Referring to FIG. 7B, in the output current IO in the circuit in FIG. 4increases, for instance by approximately a factor of two, the ripple ofthe intermediate voltage VX increases, and the minimum intermediatevoltage VMIN decreases and therefore for a constant output voltage VOthe margin (i.e. across inductor 316) in the regulator 320 decreases.However, if the voltage margin decreases below a threshold (greater thanzero), the operation of the regulator 320 is impeded.

Referring to FIG. 7C, to provide the regulator 320 with a sufficientvoltage margin voltage the switching frequency can be increases (andcycle time decreased), for example, to restore the margin shown in FIG.7A. Generally, in this example, doubling the switching frequencycompensates for the doubling of the output current IO. However moregenerally, such direct relationships between output current IO or othersensed signals and switching frequency are not necessary.

In general, a number of embodiments adapt the switching frequency of thecharge pump 100 or determine the specific switching time instants basedon measurements within the charge pump 100 and optionally in thelow-voltage and/or high-voltage peripherals coupled to the terminals ofthe charge pump 100.

In a feedback arrangement shown in FIG. 4, the controller 350 adapts(e.g., in a closed loop or open loop arrangement) the switchingfrequency. For any current up to a maximum rated current with a fixedswitching frequency, the charge pump 100 generally operates at aswitching frequency lower than (i.e., switching times greater than) aparticular minimum frequency determined by that maximum rated current.Therefore, when the current is below the maximum, capacitive losses maybe reduced as compared to operating the charge pump 100 at the minimumswitching frequency determined by the maximum rated current.

One approach to implementing this feedback operation is to monitor theintermediate voltage VX and adapt operation of the charge pump tomaintain VMIN above a fixed minimum threshold. One way to adapt theoperation of the charge pump 100 is to adapt a frequency for theswitching of the charge pump 100 in a feedback configuration such thatas the minimum intermediate voltage VMIN approaches the threshold, theswitching frequency is increased, and as it rises above the thresholdthe switching frequency is reduced. One way to set the fixed minimumthreshold voltage is as the maximum (e.g., rated) output voltage VO ofthe regulator 320, plus a minimum desired margin above that voltage. Asintroduced above, the minimum margin (greater than zero) is required toallow a sufficient voltage differential (VX−VO) to charge (i.e.,increase its current and thereby store energy in) the inductor 326 at areasonable rate. The minimum margin is also related to a guarantee on amaximum duty cycle of the regulator 320.

A second approach adapts to the desired output voltage VO of theregulator 320. For example, the regulator 320 may have a maximum outputvoltage VO rating equal to 3.3 volts. With a desired minimum margin of0.7 volts, the switching of the charge pump 100 would be controlled tokeep the intermediate voltage VX above 4.0 volts. However, if theconverter is actually being operated with an output voltage VO of1.2volts, then the switching frequency of the charge pump 100 can bereduced to the point that the intermediate voltage VX falls as low as1.9 volts and still maintain the desired margin of 0.7 volts.

In a variant of the second approach, rather than monitoring the actualoutput voltage VO, an average of the voltage between the switches 312,314 may be used as an estimate of the output voltage VO.

In yet another variant, the switching frequency of the charge pump 100is adapted to maintain the intermediate voltage VX below a thresholdvalue. For example, the threshold can be set such that the intermediatevoltage VX lowers or rises a specific percentage below or above theaverage of the intermediate voltage VX (e.g. 10%). This threshold wouldtrack the intermediate voltage VX. Similarly, a ripple relative to anabsolute ripple voltage (e.g. 100 mV) can be used to determine theswitching frequency.

Note also that the voltage ripple on the output voltage VO depends (notnecessarily linearly) on the voltage ripple on the intermediate voltageVX, and in some examples the switching frequency of the charge pump 100is increased to reduced the ripple on the output voltage VO to a desiredvalue.

Other examples measure variation in internal voltages in the charge pump100, for example, measuring the ripple (e.g., absolute or relative tothe maximum or average) across any of the capacitors C1 through C4. Suchripple values can be used instead of using the ripple on theintermediate voltage VX in controlling the switching frequency of thecharge pump 100. Other internal voltages and/or currents can be used,for example, voltages across switches or other circuit elements (e.g.,transistor switches), and the switching frequency can be adjusted toavoid exceeding rated voltages across the circuit elements.

In addition to the desired and/or actual output voltages or currents ofthe regulator 320 being provided as a control input to the controller350, which adapts the switching frequency of the charge pump 100, othercontrol inputs can also be used. One such alternative is to measure theduty cycle of the regulator 320. Note that variation in the intermediatevoltage VX affects variation in current in the Buck converter's inductor326. For example, the average of the intermediate voltage VX isgenerally reduced downward with reducing of the switching frequency ofthe charge pump 100. With the reduction of the average output voltageVO, the duty cycle of the regulator 320 generally increases to maintainthe desired output voltage VO. Increasing the duty cycle generallyincreases the efficiency of a Buck converter. So reducing the switchingfrequency of the charge pump 100 can increase the efficiency of theregulator 320.

It should be understood that although the various signals used tocontrol the switching frequency may be described above separately, theswitch frequency can be controlled according to a combination ofmultiple of the signals (e.g., a linear combination, nonlinearcombination using maximum and minimum functions, etc.). In someexamples, an approximation of an efficiency of the charge pump isoptimized.

The discussion above focuses on using the controller 350 to adjust theswitching frequency of the charge pump 100 in relatively slow scalefeedback arrangement. The various signals described above as inputs tothe controller 350 can be used on an asynchronous operating mode inwhich the times at which the charge pump 100 switches between cycles isdetermined according to the measurements. As one example, during stateone as illustrated in FIG. 6, the intermediate voltage VX falls, andwhen VX−VO reaches a threshold value (e.g., 0.7 volts), the switches inthe charge pump 100 are switched together from state one to state two.Upon the transition to state two, the intermediate voltage VX rises andthen again begins to fall, and when VX−VO again reaches the thresholdvalue, the switches in the charge pump 100 are switched together fromstate two back to state one.

In some examples, a combination of asynchronous switching as well aslimits or control on average switching frequency for the charge pump areused.

Unfortunately, as the intermediate current IX decreases the switchingfrequency of the charge pump 100 decreases as well. This can beproblematic at low currents because the frequency could drop below 20kHz, which is the audible limit for human hearing. Therefore, once thefrequency has dropped below a certain limit, a switch 344 closes andintroduces a compensation capacitor 342. This forces the converter intonon-adiabatic operation allowing the frequency to be fixed to a lowerbound (e.g. 20 kHz). Consequently, the compensation capacitor 342 isintroduced when either the duty cycle is low or when the output currentIO is low.

Note that the examples above concentrate on a compensation circuit thatpermits selectively switching a compensation capacitor of a certainfixed capacitance onto the output of the charge pump. More generally, awide variety of compensation circuits can be controlled. One example isa variable capacitor, which can be implemented as a switched capacitorbank, for example, with power of two capacitances. The optimal choice ofcapacitance generally depends on the combination of operating conditions(e.g., average current, pulsed current duty cycle, etc.) and/or circuitconfigurations (e.g., type of regulators, sources, load, pumpcapacitors), with the determining of the desired capacitance being basedon prior simulation or measurement or based on a mechanism that adjuststhe capacitance, for instance, in a feedback arrangement. In addition,other forms of compensation circuits, for example, introducinginductance on the output path, networks of elements (e.g., capacitors,inductors).

Note that the description focuses on a specific example of a chargepump. Many other configurations of charge pumps, including Dickson pumpswith additional stages or parallel phases, and other configurations ofcharge pumps (e.g., series-parallel), can be controlled according to thesame approach. In addition, the peripherals at the high and/or lowvoltage terminals are not necessarily regulators, or necessarilymaintain substantially constant current. Furthermore, the approachesdescribed are applicable to configurations in which a high voltagesupply provides energy to a low voltage load, or in which a low voltagesupply provides energy to a high voltage load, or bidirectionalconfigurations in which energy may flow in either direction between thehigh and the low voltage terminal of the charge pump. It should also beunderstood that the switching elements can be implemented in a varietyof ways, including using Field Effect Transistors (FETs) or diodes, andthe capacitors may be integrated into a monolithic device with theswitch elements and/or may be external using discrete components.Similarly, at least some of the regulator circuit may in some examplesbe integrated with some or all of the charge pump in an integrateddevice.

Implementations of the approaches described above may be integrated intoan integrated circuit that includes the switching transistors of thecharge pump, either with discrete/off-chip capacitors or integratedcapacitors. In other implementations, the controller that determines theswitching frequency of the charge pump and/or the compensation circuitmay be implemented in a different device than the charge pump. Thecontroller can use application specific circuitry, a programmableprocessor/controller, or both. In the programmable case, theimplementation may include software, stored in a tangible machinedreadable medium (e.g., ROM, etc.) that includes instructions forimplementing the control procedures described above.

It is to be understood that the foregoing description is intended toillustrate and not to limit the scope of the invention, which is definedby the scope of the appended claims. Other embodiments are within thescope of the following claims.

1-26 (canceled)
 27. An apparatus comprising a switched-capacitor chargepump configured to provide voltage conversion between a high-voltageterminal and a low-voltage terminal thereof, a compensation circuitcoupled to a first terminal of said charge pump, said first terminalbeing one of said high-voltage terminal and said low-voltage terminal,said compensation circuit having a first configuration in which saidfirst terminal of said charge pump couples to a capacitance and a secondconfiguration in which said first terminal of said is coupled to saidfirst terminal of said charge pump, and wherein, in said secondconfiguration, said capacitance is decoupled from said first terminal ofsaid charge pump, and a controller coupled to said charge pump and saidcompensation circuit, said controller comprising an output forconfiguring said compensation circuit, and a first sensor input foraccepting a first sensor signal that, at least in part, characterizesoperation of a circuit selected from the group consisting of said chargepump and a peripheral circuit directly coupled to said charge pump,wherein said controller is configured to configure said compensationcircuit based at least in part on said sensor signal to promoteefficiency of power conversion between a power source coupled to saidcharge pump and said load coupled to said charge pump via saidcompensation circuit.
 28. The apparatus of claim 27, wherein said firstsensor signal characterizes a voltage at said peripheral circuit. 29.The apparatus of claim 27, further comprising a regulator, wherein saidregulator is coupled to said compensation circuit.
 30. The apparatus ofclaim 27, wherein said charge pump comprises a capacitor, and whereinsaid first sensor signal characterizes a voltage across said capacitor.31. The apparatus of claim 27, wherein said current passing to or fromsaid charge pump via said compensation circuit comprises a currentpassing between said compensation circuit and a peripheral circuitcoupled to said charge pump via said compensation circuit.
 32. Theapparatus of claim 27, wherein said compensation circuit is coupled to aregulator that achieves a selected output voltage by adjustment of aduty cycle thereof.
 33. The apparatus of claim 27, further comprising aregulator, wherein said regulator is coupled between said compensationcircuit and said high-voltage terminal.
 34. The apparatus of claim 27,wherein said controller comprises a second sensor input for accepting asecond sensor signal that, at least in part, characterizes operation ofsaid circuit.
 35. The apparatus of claim 27, further comprising aregulator, wherein said regulator is coupled between said compensationcircuit and said low-voltage terminal.
 36. The apparatus of claim 27,wherein said controller is configured to determine an operating mode atleast in part based on said first sensor signal, and to determine saidconfiguration of said compensation circuit according to said determinedmode.
 37. The apparatus of claim 27, wherein said first sensor signalcharacterizes a voltage at a low-voltage terminal.
 38. The apparatus ofclaim 27, wherein said controller is configured to couple said firstterminal to a selected capacitance at times that optimizepower-conversion efficiency.
 39. The apparatus of claim 27, wherein saidfirst sensor signal characterizes said switching frequency of saidcharge pump.
 40. The apparatus of claim 27, wherein said first sensorsignal characterizes a voltage at a high-voltage terminal.
 41. Theapparatus of claim 27, wherein said charge pump comprises a Dicksoncharge pump.
 42. The apparatus of claim 27, wherein said first sensorsignal characterizes a duty cycle of a pulsed current passing to or fromsaid charge pump.
 43. The apparatus of claim 27, wherein said firstsensor signal characterizes an average of a current passing to saidcharge pump.
 44. A method comprising carrying out voltage conversion,wherein carrying out voltage conversion comprises receiving a sensorsignal that characterizes, at least in part, operation of a circuit thatis selected from the group consisting of a switched-capacitor chargepump that provides voltage conversion between terminals that include ahigh-voltage terminal and a low-voltage terminal and a peripheralcircuit that is directly connected to a switched capacitor charge pumpthat provides voltage conversion between terminals that include ahigh-voltage terminal and a low-voltage terminal, and based at least inpart on said sensor signal, causing a compensation circuit that iscoupled to a first terminal of said charge pump to transition betweencoupling and decoupling a capacitance from said first terminal, whereinsaid first terminal is selected from said group consisting of ahigh-voltage terminal of said charge pump and a low-voltage terminal ofsaid charge pump, wherein causing said compensating circuit totransition comprises causing said compensation circuit to transition attimes that promote efficiency of power conversion between a power sourcecoupled to said charge pump and a load coupled to said charge pump viasaid compensation circuit.
 45. An apparatus comprising a powerconverter, said power converter first and second terminals, said powerconverter being configured to cause a second voltage to be maintained atsaid second terminal in response to presence of a first voltagepresented at said first terminal, wherein said power converter comprisesa compensation circuit, a controller, a switching network, and acapacitors, wherein said switching network interconnects saidcapacitors, wherein, as a result of transitioning between first andsecond states thereof, said switching network causes said capacitors totransition between corresponding first and second arrangements, whereinas a result of a transition, electrical charge propagates between saidcapacitors, wherein said controller is connected to receive, from atleast one of a first circuit and a second circuit, informationindicative of an extent to which said propagation of said electricalcharge between said capacitors results in energy loss, wherein saidcontroller is configured to cause said compensation circuit totransition between a first configuration and a second configurationbased on said information, said transition being one that reduces saidextent and that causes a capacitance of said compensation circuit to beswitched into or out of communication with said first circuit, whereinsaid first circuit is a circuit that is formed by said switching networkand said capacitors, and wherein said second circuit is a circuit thatis directly connected to a circuit that is formed by said capacitors andsaid switching network.
 46. The apparatus of claim 45, wherein saidsecond circuit comprises an inductor.